The present invention relates to establishment of the complex measuring capacity of measuring apparatus with mixer detection.
In ac signal processing systems, it is often necessary to determine quantities to be measured, for example currents or voltages, as to their magnitude and phase. The ability to test signals as to their magnitude and phase is called complex measuring capacity or vectorial measuring capacity.
It is usually necessary to measure such values in order to determine accurately the properties of electrical circuits. Devices intended for measuring these parameters are called network analyzers. They can be assigned to two groups, namely scalar and vectorial network analyzers, depending on whether the values of interest, i.e., the network parameters, are to be determined as to their frequency only or as to their magnitude and phase.
With higher frequencies, it becomes increasingly more difficult to determine the respective parameters of the signals. As a rule, the signal to be measured is transformed into lower frequency oscillations by a heterodyne mixing process. The lower frequency oscillations are proportional in amplitude and phase to the high frequency signal, i.e., they contain all of the information of interest.
Since such apparatus operating on the basis of the heterodyne principle entails considerable costs, less costly apparatus operating on the homodyne principle is substituted for it whenever possible. Such apparatus operates with a single standard signal generator and is very sturdy. However, a disadvantage of this detection process is that it only provides real information. In general, two operations are common. First, measurement of amplitude, which is performed by means of, for example, detector diodes or output meters. [4][12] Secondly, detection by means of mixers, which provide information proportional to the real part of the complex signal to be measured. [1] In the case of scalar-network analyzers it is customary to work only with the information regarding the frequency of the test signals. This apparatus will not be further discussed here.
The disadvantage of test signals with real values is that they must be compensated for in order to provide vectorial network analyzers. For this purpose, interference principles, in which two different signals are superposed in a different manner, are often used in the detection of amplitudes. With a sufficient number of superpositions it then is possible to provide conclusions as to the ratio of the two signals or of a value proportional thereto. In contrast to this, in connection with mixer detection the signal to be tested is measured once in a direct manner and then in a phase-shifted manner.
FIGS. 1-4 illustrate the detection part of homodyne systems. The system shown in each of these figures can be employed, in subsequent figures, as a device which will be identified as a KHD.
The device in accordance with FIG. 1 is used to explain the detecting process. The amplitudes of the signal a to be measured and present at the mixer 1 and of the local oscillator signal b are selected in such a way that the mixer operates linearly, i.e., EQU .vertline.a.vertline.&lt;&lt;.vertline.b.vertline..
In the steady state of signal a, the mixer output dc voltage U, filtered out by the low pass filter 2, is ##EQU1## where .rarw. is the real mixer conversion efficiency. By means of a suitable choice of the planes of reference it is always possible to make EQU U=.eta.RE a (2)
Furthermore, it is possible to assume the mixer conversion efficiency to be 1 (.eta.=1), because it can be mentally transposed into the calibrating values, which are also not known and need to be determined by appropriate measurements.
The fact that the complex measuring capability is unknown--U in Eq. (2) is only a real value--can be remedied in a number of ways. In this connection, a few explanatory initial considerations with reference FIG. 2 follow. In FIG. 2, the signal a to be detected is split into two portions in the signal splitter 3, where EQU a.sub.1 =a, a.sub.2 =k a (3)
applies. In a manner analogous to Eq. (2), the dc voltages filtered out by the low-pass filters 5 and 6 then are EQU U.sub.1 =Re a, U.sub.2 =Re(ka) (4)
It is easy to show by insertion that by means of EQU U(p)=U.sub.1 +p U.sub.2 ( 5)
a fictitious value is formed which is proportional to a and thus has a complex value, if ##EQU2## applies. Here, k is the effective phase displacement generated by the phase shifter 7. A particularly advantageous case is k=e.sup.-j90.degree., so that p=j and U(p) =U.sub.1 +jU.sub.2. But also the general case, that .vertline.k.vertline..noteq.1 and arc(k).noteq..+-.90.degree., are included with Eqs. (5) and (6), in which case Im k.noteq.0 is required.
A problem arises from the fact that the characteristic of the phase shifter is generally not known a priori, so that the effective characteristic of the phase shifter or, because of Eq. (6), similarly the complex weighting factor p must be determined in an appropriate manner by measuring. The procedure of assuming an approximate value for k while accepting possible errors is common. Such an approximate value can be provided by external measurements performed on the phase shifter. This is an approximate value for the reason that because of the coupling of the phase shifter with its wiring--for example resulting in multiple reflections--a different phase shift value becomes effective in connection with the actual test set-up than with the external measuring apparatus. It is therefore necessary to determine the effective characteristic k of the phase shifter in situ. In accordance with the state of the art this can be accomplished if, for example, additional phase shifts are generated by the addition of unknown binary phase shifters, so that sufficient measured values are available to determine the value k of interest or also, simultaneously, p. [9] The methods according to the present invention are distinguished by broader applicability, a smaller number of additionally required components, clearly reduced demands on the latter and clearly reduced demands regarding their combination. Furthermore, the methods are not limited to a complex voltmeter, but relate to complete network analyzers which mostly have several measurement points. In accordance with the use of the method, complex measurability is then established simultaneously at all measurement points.
To explain these methods, the device will first be more generalized, for which purpose the nodes 3 and 4 of the local oscillator output distribution and the phase shifter 7 in FIG. 2 will be considered in more detail. The functions of node 3 and phase shifter 7 can often be taken over by a component 8 which, in the high frequency range, can be, for example, a coupler. It should be noted that it is also permissible to interchange parts 8 and 4, which does not, however, result in basic differences, so that hereafter only one variation will be described.
In whichever way the combination is selected, it is possible to form a complex replacement measuring value U(p) analogous to Eq. (5), which is proportional in amplitude and phase to the actual measuring value a, EQU a.about.U.sub.1 +p U.sub.2 =U(p), (7)
where a possibly present proportionality factor is treated as already described. The weighting factor p is, as already mentioned, not known a priori, so that the replacement measuring value U(p), which thus can only be formed formally, must always be noted in relation to this weighting factor.
In a variation of the previous wiring it is possible to use, instead of the signal splitting at 8 and detection with the mixers 9 and 10, a detection point which does not operate temporally parallel but serially.
In this connection consider the device shown in FIG. 3. The signal a to be detected--or here again the local oscillator signal b--is fed to a mixer 12 via a switchable binary phase shifter (BPS) 11, which can assume either one of two states in a reproducible manner. In this instance phase shifter 11 has the transmission factor c.sub.o in the first state and kc.sub.o in the second. Here, too, .vertline.k.vertline..noteq.1 is permissible, which can be caused, for example, by a parasitic amplitude modulation of the binary phase shifter. However, coupling of the binary phase shifter with other time-variable elements of the measuring apparatus is not permissible, which constitutes a certain disadvantage of this variation.
The low-pass filtered mixer output voltage in the first state of the phase shifter is U.sub.1 and, after switching, in the second is U.sub.2. Here again, ##EQU3## applies.
Besides the cases mentioned so far, the mathematical representation used can also be applied for basically different devices, so that they can be adapted to the concept of the invention. In this connection, reference is made to FIG. 4, in which either the signal a to be detected or the local oscillator signal b is fed to mixer Eq. (14), shifted by the frequency f.sub.ZF by means of a single sideband offset device (ESB). The output signal of mixer 14 is filtered by a band pass filter 15 with the center frequency f.sub.ZF, so that then an intermediate frequency signal is available, which ideally--as in a heterodyne mixing process--is proportional in amplitude and phase to the signal to be detected. The complex voltage measured is designated with U.sub.ZF, for which EQU U.sub.ZF .about.a (9)
applies but which, in the actual case, is disturbed because of the now finally satisfactory suppression of the reflecting band. This is expressed by there being an interference term proportional to the conjugated complex signal EQU U.sub.ZF .about.a+s.sub.e a* (10)
which is smaller to the degree that the complex distortion factor s.sub.e of the single sideband offset device is smaller. By means of algebraic transformations ##EQU4## it is found that this manner of detection can also be formally treated in the same way as the others previously described.
It is known to establish the complex measuring capacity of such devices by using phase shifting devices having a known phase shift value and a known parasitic amplitude modulation. Alternatively, a cascade of three binary phase shifters having unknown characteristics may be used. Each of the phase shifters is switchable between an on and an off state and by measuring an output signal associated with each of the possible combination of phase shifter states, it is possible to obtain three phase shift values and three amplitude factors. With this arrangement, the three phase shifters must be electrically decoupled from one another and they must operate in a reproducible manner.